The present invention relates to a switching power supply circuit including a voltage resonance type converter.
A current resonance type switching power source and a voltage resonance type switching power source are generally known as the so-called soft-switching power source adopting a resonance type. In the present circumstances, a current resonance type converter using a half-bridge coupling system having two switching elements is widely adopted with easiness of its practical application as a background.
However, at the present time, for example, the characteristics of a high withstanding voltage switching element have been improved. Thus, a problem about a withstand voltage when the voltage resonance type converter is put to practical use has been cleared. In addition, it is also known that the voltage resonance type converter structured using a single ended system having one switching element is advantageous in terms of an input feedback noise, a noise component on a DC output voltage line, and the like as compared with a current resonance type forward converter including one switching element.
FIG. 21 shows one structural example of a switching power supply circuit including the voltage resonance type converter using the single ended system. This switching power supply circuit is disclosed in Japanese Patent Laid-open 2000-134925.
In the switching power supply circuit shown in FIG. 21, an AC voltage from a commercial AC power source AC is rectified and smoothed by a rectifying and smoothing circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci. As a result, a rectified and smoothed voltage Ei is generated as a voltage developed across the smoothing capacitor Ci.
At that, a noise filter is provided for the line of the commercial AC power source AC. The noise filter includes a set of common mode choke coils CMC, and two across capacitors CL and serves to remove a common mode noise.
The rectified and smoothed voltage Ei is inputted as a DC input voltage to a voltage resonance type converter. The voltage resonance type converter, as described above, adopts a structure using the single ended system including one switching element Q1. In addition, the voltage resonance type converter in this case is one using a separate excitation system in which the switching element Q1 constituted by a MOS-FET is switching-driven by an oscillation/drive circuit 2.
A body diode DD of the MOS-FET is connected in parallel with the witching element Q1. In addition, a primary side parallel resonance capacitor Cr is connected in parallel between a drain and a source of the switching element Q1.
The primary side parallel resonance capacitor Cr forms, together with a leakage inductance L1 of a primary winding N1 of an insulating converter transformer PIT, a primary side parallel resonance circuit (voltage resonance circuit). A voltage resonance type operation as a switching operation of the switching element Q1 is obtained by the primary side parallel resonance circuit.
The oscillation/drive circuit 2 applies a gate voltage as a drive signal to a gate of the switching element Q1 to switching-drive the switching element Q1. As a result, the switching element Q1 performs a switching operation at a switching frequency corresponding to a period of the drive signal.
The insulating converter transformer PIT transmits a switching output from the switching element Q1 to its secondary side.
The insulating converter transformer PIT, for example, includes an EE letter-like core in which two E letter-like cores made of a ferrite material are combined with each other in terms of a structure thereof. After a winding portion is divided between the primary side and the secondary side, the primary winding N1 and a secondary winding N2 are wound around a central magnetic leg of the EE letter-like core.
In this case, a gap having a gap length of about 1.0 mm is formed in the central magnetic leg of the EE letter-like core of the insulating converter transformer PIT. As a result, a coupling coefficient k of about 0.8 to about 0.85 is obtained between the primary side and the secondary side. A saturated state becomes difficult to obtain all the more because the coupling coefficient k of this degree is regarded as loose coupling in terms of the degree of coupling. In addition, a value of the coupling coefficient k becomes a setting factor for a leakage inductance (L1).
One end of the primary winding N1 of the insulating converter transformer PIT is inserted between the switching element Q1 and a positive-polarity terminal of the smoothing capacitor Ci, which results in that the switching output from the switching element Q1 is transmitted. An AC voltage which is induced by the primary winding N1 is generated in the secondary winding N2 of the insulating converter transformer PIT.
In this case, a secondary side series resonance capacitor C2 is connected in series with one end of the secondary winding N2, whereby a leakage inductance L2 of the secondary winding N2 and a capacitance of the secondary side series resonance capacitor C2 form a secondary side series resonance circuit (current resonance circuit).
In addition, rectifying diodes Do1 and Do2, and a smoothing capacitor Co are connected to a secondary side series resonance circuit as shown in FIG. 21, thereby structuring a voltage doubler half-wave rectifying circuit. The voltage doubler half-wave rectifying circuit generates a secondary side DC output voltage Eo at a level which is double as large as that of an AC voltage V2 induced in the secondary winding N2 in terms of a voltage developed across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied to a load and is inputted as a detection voltage for a constant voltage control to a control circuit 1.
The control circuit 1 inputs a detection output which is obtained by detecting the level of the secondary side DC output voltage Eo inputted thereto as the detection voltage to the oscillation/drive circuit 2.
The oscillation/drive circuit 2 outputs a drive signal a frequency of which is made variable in correspondence to the level of the secondary side DC output voltage Eo indicated by the detection signal inputted thereto. Thus, the oscillation/drive circuit 2 controls the switching operation of the switching element Q1 so that the secondary side DC output voltage Eo becomes constant at a predetermined level. As a result, the stabilization control for the secondary side DC output voltage Eo is carried out.
FIGS. 22A, 22B and 22C, and FIG. 23 show results of experiments on the power supply circuit having the structure shown in FIG. 21. At that, when the experiments were performed, the setting was made for the main portions of the power supply circuit shown in FIG. 21 as follows.
In the insulating converter transformer PIT, EER-35 was selected for a core, and a gap having a gap length of 1 mm was set in the central magnetic leg. In addition, the number T of turns of the primary winding N1, and the number T of turns of the secondary winding N2 were set to 39T and 23T, respectively. Also, an induced voltage level per one turn (T) of the secondary winding N2 was set to 3V/T. The coupling coefficient k of the insulating converter transformer PIT was set to 0.81.
In addition, a capacitance of the primary side parallel resonance capacitor Cr was selected as 3,900 pF, and a capacitance of the secondary side series resonance capacitor C2 was selected as 0.1 μF. As a result, a resonance frequency fo1 of the primary side parallel resonance circuit, and a resonance frequency fo2 of the secondary side series resonance circuit were set as 230 kHz and 82 kHz, respectively. In this case, a relative relationship between the resonance frequencies fo1 and fo2 can be expressed by fo1≈2.8×fo2.
A rated level of the secondary side DC output voltage Eo is 135 V. A corresponding load power ranges from a maximum load power Pomax=200 W to a minimum load power Pomin=0 W.
FIGS. 22A, 22B and 22C are waveform charts showing operations of the main portions in the power supply circuit shown in FIG. 21 based on a switching period of the switching element Q1. That is to say, FIG. 22A shows waveforms of a voltage V1, a switching current IQ1, a primary winding current I1, a secondary winding current I2, and secondary side rectified currents ID1 and ID2 during a phase of the maximum load power Pomax=200 W. FIG. 22B shows waveforms of the voltage V1, the switching current IQ1, the primary winding current I1, and the secondary winding current I2 during a phase of an intermediate load power Po=120 W. Also, FIG. 22C shows waveforms of the voltage V1 and the switching current IQ1 during a phase of the minimum load power Pomin=0 W.
The voltage V1 is one that is developed across the switching element Q1. That is to say, the voltage V1 has a waveform in which the voltage V1 is set at a 0 level during an ON-time period TON for which the switching element Q1 is in an ON state, and turns into a sine-wave resonance pulse for an OFF-time period TOFF for which the switching element Q1 is in an OFF state. Thus, the resonance pulse waveform of the voltage V1 shows that the operation of the primary side switching converter is of the voltage resonance type.
The switching current IQ1 is a current which is caused to flow through the switching element Q1 (and the body diode DD). That is to say, the switching current IQ1 is caused to flow during the ON-time period TON so as to show the waveform of e.g., FIG. 22A, and is at a 0 level during the OFF-time period TOFF.
The primary winding current I1 which is caused to flow through the primary winding N1 is obtained by composing a current component which is caused to flow as the above-mentioned switching current IQ1 during the ON-time period TON, and a current which is caused to flow through the primary side parallel resonance capacitor Cr during, the OFF-time period TOFF with each other.
In addition, while illustrated only in FIG. 22A, the rectified currents ID1 and ID2 which show the operation of the secondary side rectifying circuit and which are caused to flow through the rectifying diodes Do1 and Do2 have sine-wave waveforms as shown in the figure, respectively. In this case, the resonance operation of the secondary side series resonance circuit dominantly appears in the waveform of the rectified current ID1 rather than in the waveform of the rectified current ID2.
The secondary winding current I2 which is caused to flow through the secondary winding N2 is obtained by composing the rectified currents Id1 and ID2 with each other.
FIG. 23 shows a switching frequency fs against load fluctuation, the ON-time period TON and OFF-time period TOFF of the switching element Q1, and an efficiency ηAC→DC of converting an AC power to a DC power in the power supply circuit shown in FIG. 21.
Firstly, looking at the efficiency ηAC→DC of converting an AC power to a DC power, it is understood that the high efficiency of 90% or more is obtained in a wide range of the load power Po from 50 W to 200 W. The inventor of this application has formerly confirmed from the experiments that such characteristics are obtained when the voltage resonance type converter using the single ended system is combined with the secondary side series resonance circuit.
In addition, the switching operation representing the constant voltage control characteristics against the load fluctuation caused in the power supply circuit shown in FIG. 21 is shown based on the switching frequency fs, the ON-time period TON and the OFF-time period TOFF of FIG. 23. In this case, the switching frequency fs is nearly constant against the load fluctuation. On the other hand, the ON-time period TON and the OFF-time period TOFF linearly change so as to show tendencies opposite to each other as shown in FIG. 23. This fact shows that after the switching frequency (switching period) is made nearly constant against the fluctuation of the secondary side DC output voltage Eo, the switching operation is controlled so that a time period ratio of the ON-time period to the OFF-time period is changed. Such control can be regarded as pulse width modulation (PWM) control for making the ON/OFF-time period within one period variable. In the power supply circuit shown in FIG. 21, the secondary side DC output voltage Eo is stabilized by the PWM control.
FIG. 24 schematically shows the constant voltage control characteristics of the power supply circuit shown in FIG. 21 based on a relationship between the switching frequency fs (kHz) and the secondary side DC output voltage Eo.
The power supply circuit shown in FIG. 21 includes the primary side parallel resonance circuit and the secondary side series resonance circuit. Hence, the power supply circuit shown in FIG. 21 compoundly has two resonance impedance characteristics, i.e., resonance impedance characteristics corresponding to a resonance frequency fo1 of the primary side parallel resonance circuit, and resonance impedance characteristics corresponding to a resonance frequency fo2 of the secondary side series resonance circuit. In addition, the power supply circuit shown in FIG. 21 has a relationship of fo1≈2.8×fo2. Hence, as shown in FIG. 24 as well, the secondary side series resonance frequency fo2 is lower than the primary side parallel resonance frequency fo1.
Then, there is supposed the constant voltage control characteristics for the switching frequency fs under the condition of a certain AC input voltage VAC. In this case, as shown in the figure, the constant voltage control characteristics during the maximum load power Pomax phase and during the minimum load power Pomin phase under the resonance impedance corresponding to the resonance frequency fo1 of the primary side parallel resonance circuit are represented by characteristic curves A and B, respectively. Also, the constant voltage control characteristics during the maximum load power Pomax phase and during the minimum load power Pomin phase under the resonance impedance corresponding to the resistance frequency fo2 of the secondary side series parallel circuit are represented by characteristic curves C and D, respectively. When the constant voltage control is intended to be performed based on a rated level tg of the secondary side DC output voltage Eo under the characteristics shown in FIG. 24, a variable range (necessary control range) of the switching frequency fs required for the constant voltage control can be expressed as a section indicated by Δfs.
The necessary control range Δfs shown in FIG. 24 ranges from the characteristic curve C during the maximum load power Pomax phase corresponding to the resonance frequency fo2 of the secondary side series resonance circuit to, the characteristic curve B during the minimum load power Pomin phase corresponding to the resonance frequency fo1 of the primary side parallel resonance circuit. Thus, the necessary control range Δfs strides across the characteristic curve D during the minimum load Pomin phase corresponding to the resonance frequency fo2 of the secondary side series resonance circuit and the characteristic curve A during the maximum load power Pomax phase corresponding to the resonance frequency fo1 of the primary side parallel resonance circuit.
For this reason, as for the constant voltage control operation of the power supply circuit shown in FIG. 21, the switching drive control is carried out based on a state of the PWM control in which the time period ratio of the ON-time period to the OFF-time period for one switching period is changed after the switching frequency fs is nearly fixed. At that, this is also shown based on a situation in which the widths of the OFF-time period TOFF and ON-time period TON change under such a condition that the time period lengths of one switching period (TOFF+TON) shown in the phase of the maximum load power Pomax=200 W, the phase of the load power Po=100 W, and the phase of the minimum load power Pomin=0 W shown in FIGS. 22A, 22B and 22C, respectively, are made nearly constant.
It is thought that such an operation is obtained by making a change between a first state and a second state under the narrow variable range (Δfs) of the switching frequency. In the first state, the resonance impedance (capacitive impedance) having the resonance frequency fo1 of the primary side parallel resonance circuit becomes dominant in terms of the resonance impedance characteristics corresponding to the load fluctuation in the power supply circuit. Also, in the second state, the resonance frequency fo2 (inductive impedance) of the secondary side series resonance circuit becomes dominant in terms of these resonance impedance characteristics.
However, the power supply circuit shown in FIG. 21 involves the problem as will be described below.
That is to say, the switching current IQ1 during the maximum load power Pomax phase shown in FIG. 22A which has been formerly described is held at the 0 level until an end point of the OFF-time period TOFF as the turn-ON timing is reached. When the ON-time period TON is reached, firstly, the negative-polarity current is caused to flow through the body diode DD, and thereafter is inversed in polarity to be caused to flow through the drain and source of the switching element Q1. In such a manner, the power supply circuit operates. This operation shows a state in which a zero voltage switching (ZVS) is properly carried out.
On the other hand, for the switching current IQ1 during the phase of the load power Po=120 W corresponding to the intermediate load, the operation is obtained in which the switching current IQ1 is caused to flow in the form of noises at the timing at and before the end point of the OFF-time period as the turn-ON timing is reached. This operation is an abnormal operation in which no ZVS is properly carried out.
That is to say, it has been found out that the voltage resonance type converter including the secondary side series resonance circuit, as shown in FIG. 21, performs the abnormal operation in which no ZVS is properly carried out during the intermediate load phase. It has been confirmed that the power supply circuit shown in FIG. 21, for example, actually performs such an abnormal operation in a region of a load fluctuation range as a section A shown in FIG. 23.
As has been formerly described, the voltage resonance type converter including the secondary side series resonance circuit essentially has a tendency to show the characteristics with which the high efficiency can be satisfactorily maintained against the load fluctuation. However, as shown in terms of the switching current IQ1 of FIG. 22B, the adequate peak current is caused to flow at the turn-ON phase of the switching element Q1. Hence, an increase in switching loss is caused, which results in that the voltage resonance type converter carries a factor of reducing the power conversion efficiency.
In addition, in any case, causing the abnormal operation as described above results in that the phase-gain characteristics of the constant voltage control circuit system, for example, are shifted, and thus the switching operation is performed in the abnormal oscillation state. For this reason, it is strongly recognized under the existing circumstances that it is actually difficult to put the voltage resonance type converter including the secondary side series resonance circuit to practical use.